Not Applicable
1. Field of the Invention
This application relates to digital implementation of electronic article surveillance (EAS) detection filtering, and more particularly to detection filtering in pulsed EAS systems.
2. Description of the Related Art
EAS systems, such as disclosed in U.S. Pat. Nos. 4,622,543, and 6,118,378 transmit an electromagnetic signal into an interrogation zone. EAS tags in the interrogation zone respond to the transmitted signal with a response signal that is detected by a corresponding EAS receiver. Previous pulsed EAS systems, such as ULTRA*MAX sold by Sensormatic Electronics Corporation, use analog electronics in the receiver to implement detection filters with either a quadrature demodulation to baseband or an envelope detection from an intermediate frequency conversion. The EAS tag response is a narrow band signal, in the region of 58000 hertz, for example.
An EAS tag behaves as a second order resonant filter with response
s(t)=Axc2x7exe2x88x92xcex1xc2x7txc2x7sin(2xc2x7xcfx80xc2x7f0xc2x7t+xcex8),
where A is the amplitude of the tag response, f0 is the natural frequency of the tag, and xcex1 is the exponential damping coefficient of the tag. The natural frequency of the tag is determined by a number of factors, including the length of the resonator and orientation of the tag in the interrogation field, and the like. Given the population of tags and possible trajectories through the interrogation zone, the natural frequency is a random variable. The probability distribution of the natural frequency has a bell shaped curve somewhat similar to Gaussian. For simplifying the receiver design it may be assumed uniform without a great loss in performance. Its distribution is assumed to be bounded between some minimum and maximum frequencies, fmin and fmax, respectively.
The exponential damping coefficient xcex1, in effect, sets the bandwidth of the tag signal. Nominal values for xcex1 are around 600 with magnetomechanical or acousto-magnetic type tags. On the other hand, for ferrite tags cc will be much larger, on the order of 1200 to 1500.
The phase of the tag response depends on the transmit signal and many of the same parameters as the natural frequency. The transmit signal determines the initial conditions on the tag when the transmitter turns off. This sets the phase of the response as it goes through its natural response. The amplitude of the tag""s response is dependent on all of the same parameters: orientation and position in the field, physics of the tag, etc.
Pulse EAS systems, such as ULTRA*MAX systems, operating around 60000 Hz preside in a low frequency atmospheric noise environment. The statistical characteristic of atmospheric noise in this region is close to Gaussian, but somewhat more impulsive, e.g., a symmetric xcex1-stable distribution with characteristic exponent near, but less than, 2.0. In addition to atmospheric noise, the 60000 hertz spectrum is filled with man made noise sources in a typical office/retail environment. These man made sources are predominantly narrow band, and almost always very non-Gaussian. When many of these sources are combined with no single dominant source, the sum approaches a normal distribution due to the Central Limit Theorem. The classical assumption of detection in additive white Gaussian noise is used herein. The xe2x80x9cwhitexe2x80x9d portion of this assumption is reasonable since the receiver input bandwidth of 3000 to 5000 hertz is much larger than the signal bandwidth. The Gaussian assumption is justified as follows.
Where atmospheric noise dominates the distribution is known to be close to Gaussian. Likewise, where there are a large number of independent interference sources the distribution is close to Gaussian due to the Central Limit Theorem. If the impulsiveness of the low frequency atmospheric noise were taken into account, then the locally optimum detector could be shown to be a matched filter preceded by a memoryless nonlinearity (for the small signal case). The optimum nonlinearity can be derived using the concept of xe2x80x9cinfluence functionsxe2x80x9d. Although this is generally very untractable, there are several simple nonlinearities that come close to it in performance. To design a robust detector some form of nonlinearity must be included.
When there is a small number of dominant noise sources we include other filtering, prior to the detection filters, to deal with these sources. For example, narrow band jamming is removed by notch filters or a reference based LMS canceller. After these noise sources have been filtered out, the remaining noise is close to Gaussian.
Referring to FIG. 1, when the signal of interest is completely known a matched filter is the optimum detector. In our case, say we knew the resonant frequency of the tag and its precise phase angle when ringing down. The signal we""re trying to detect is
s(t)=Axc2x7exe2x88x92xcex1xc2x7txc2x7sin(2xc2x7xcfx80xc2x7f0xc2x7t+xcex8).
Then the matched filter is simply the time reversed (and delayed for causality) signal, s(Trxe2x88x92t) at 2. The matched filter output is sampled at 4 at the end of the receive window, Tr, and compared to the threshold at 6. A decision signal can be sent depending on the results of the comparison to the threshold. The decision can be a signal to sound an alarm or to take some other action. Note that we do not have to know the amplitude, A. This is because the matched filter is a xe2x80x9cuniformly most powerful testxe2x80x9d with regard to this parameter. This comment applies to all the variations of matched filters discussed below.
Referring to FIG. 2, when the signal of interest is completely known except for its phase xcex8, then the optimum detector is the quadrature matched filter (QMF). QMF is also known as noncoherent detection, since the receiver is not phase coherent with the received signal. On the other hand, the matched filter is a coherent detector, since the phase of the receiver is coherent with the received signal. The receive signal r(t) which includes noise and the desired signal s(t) is filtered by s(Trxe2x88x92t) at 8 as in the matched filter, and again slightly shifted in phase by xcfx80/2 at 10. The outputs of 8 and 10 are each squared at 12, combined at 14, sampled at 16, and compared to the threshold at 18.
Referring to FIG. 3, when the signal of interest is completely known except for its frequency fn and phase xcex8, then the optimum detector is a bank of quadrature matched filters (QMFB). A quadrature matched filter bank can be implemented as a plurality of quadrature matched filters 20, 22, 24, and 26, which correlate to quadrature matched filters with center frequencies of f1, f2 through fn, respectively. The outputs of the quadrature matched filters are summed at 28, sampled at 29 and compared to a threshold at 30.
Referring to FIG. 4 a block diagram of a conventional analog EAS receiver is illustrated. The antenna signal 32 passes through a gain and filtering stage 34 with center frequency equal to the nominal tag frequency and bandwidth of about 3000 hertz, for example. Following this, the signal is demodulated to baseband with a quadrature local receive oscillator 36. The oscillator frequency may or may not be matched precisely to the transmit frequency. Furthermore, the oscillator phase is not necessarily locked to the transmit oscillator""s phase.
The in-phase (I) and quadrature-phase (Q) baseband components are subsequently lowpass filtered by the in-phase 38 and quadrature-phase 40 baseband filters, respectively. This serves to remove the double frequency components produced by the mixing process, as well as further reduces the detection bandwidth. These baseband filters are typically 4th order analog filters, e.g., Butterworth and Chebychev type.
The outputs of the baseband filters 38, and 40 are passed through rectifiers 42 and 44, respectively, which removes the sign information from the I and Q components. The outputs of the rectifiers, are sampled by ADC 46 and 48, respectively, at the end of the receive window and passed into the microprocessor, where the I and Q components are squared and summed together to produce a noncoherent detection statistic.
Referring to FIG. 5, a block diagram of an alternate analog EAS receiver is illustrated. The antenna signal 50 passes through a gain and filtering stage 52 with center frequency equal to the nominal tag frequency and bandwidth of about 5000 hertz, for example. Following this, the signal is modulated to an intermediate frequency (IF) of approximately 10000 hertz with a local receive oscillator at 52. The IF signal is filtered by an IF bandpass filter 54 with bandwidth of approximately 3000 hertz to remove off frequency products from the mixer and further reduce bandwidth for the detector.
The filtered IF signal then passes through an envelope detector, which in this case is the combination of a rectifier 55 and lowpass filter 56. The output of the envelope detector is sampled by an ADC 58 and passed to the processor for detection processing. Note that envelope detection removes the phase of the receive signal. In fact, it can be shown that envelope detection is simply a different implementation of a quadrature detector, and thus it is noncoherent.
The problem presented was to design a cost-effective system, which would more reliably detect a tag response in the presence of noise. The noise environment is assumed to be close to Gaussian with much wider bandwidth than the tag signal. Some environments may include narrow band interference from electronic equipment.
The present invention provides, in a first aspect, a system and method, using a quadrature matched filter bank, to digitally detect a signal from an electronic article surveillance tag. The system and method including: filtering using a detection filter pair comprised of h(T0xe2x88x92t)xc2x7sin(xcfx89xc2x7t) and h(T0xe2x88x92t)xc2x7cos(xcfx89xc2x7t), where the envelope h(T0xe2x88x92t) contains preselected time and frequency domain properties according to the signal to be detected; squaring the output of each of the filters; summing the squared outputs of each of the filter pairs to provide a test statistic for detection of the tag signal.
The system and method further including a plurality of the filter pairs wherein each pair is at a frequency xcfx89n for 1xe2x89xa6nxe2x89xa6N, where N is selected to cover the range of uncertainty of the signal to be detected, and summing each of the squared and summed results of each of the filter pairs to provide the test statistic for detection of the tag signal. Each of the filter pairs can be matched to the response signal from the electronic article surveillance tag wherein the envelope h(T0xe2x88x92t) is the time reversed version of the signal to be detected.
In a second aspect, a system and method, using a quadrature matched filter bank with envelope estimation, for detecting the signal from an electronic article surveillance tag. The system and method including: filtering using a filter comprised of h(T0xe2x88x92t)xc2x7sin(xcfx89nxc2x7t) wherein the envelope h(T0xe2x88x92t) contains preselected time and frequency domain properties according to the signal to be detected; envelope detecting of the output of the filter; and, squaring the output of the envelope detection to provide a test statistic for detection of the tag signal.
The system and method further including a plurality of the filters wherein each filter is at a frequency xcfx89n for 1xe2x89xa6nxe2x89xa6N, where N is selected to cover the range of uncertainty of the signal to be detected; and, then summing the squared output of the plurality of filters to provide the test statistic for detection of the tag signal. Each of the filters can be matched to the response signal from the electronic article surveillance tag wherein the envelope h(T0xe2x88x92t) is the time reversed version of the signal to be detected.
In a third aspect, a system and method, using a bank of correlation receivers, for detecting a signal from an electronic article surveillance tag. The system and method including: a correlation receiver that mixes a received signal with an envelope h(t) and a pair of local oscillators cos(xcfx89xc2x7t) and sin(xcfx89xc2x7t); integrating the mixed signal over the sampling period T0; squaring the integrated output; summing the squared output for each of the pair of local oscillators to provide a test statistic for detection of the tag signal.
The system and method further including a plurality of the correlation receivers where the local oscillators cos(xcfx89nxc2x7t) and sin(xcfx89nxc2x7t) are at frequency xcfx89n for 1 less than n less than N, where N is selected to cover the range of uncertainty of the signal to be detected; and, summing the output of the plurality of correlation receivers to provide the test statistic for detection of the tag signal.
In a fourth aspect, the system and method of the third aspect where the local oscillators and the integration comprise a discrete Fourier transform
Objectives, advantages, and applications of the present invention will be made apparent by the following detailed description of embodiments of the invention.